Pole splitting is a phenomenon exploited in some forms of frequency compensation used in an electronic amplifier. When a capacitor is introduced between the input and output sides of the amplifier with the intention of moving the pole lowest in frequency (usually an input pole) to lower frequencies, pole splitting causes the pole next in frequency (usually an output pole) to move to a higher frequency. This pole movement increases the stability of the amplifier and improves its step response at the cost of decreased speed.
This example shows that introducing capacitor C<sub>C</sub> in the amplifier of Figure 1 has two results: firstly, it causes the lowest frequency pole of the amplifier to move still lower in frequency and secondly, it causes the higher pole to move higher in frequency. This amplifier has a low frequency pole due to the added input resistance R<sub>i</sub> and capacitance C<sub>i</sub>, with the time constant C<sub>i</sub> ( R<sub>A</sub> || R<sub>i</sub> ). This pole is lowered in frequency by the Miller effect. The amplifier is given a high frequency output pole by addition of the load resistance R<sub>L</sub> and capacitance C<sub>L</sub>, with the time constant C<sub>L</sub> ( R<sub>o</sub> || R<sub>L</sub> ). The upward movement of the high-frequency pole occurs because the Miller-amplified compensation capacitor C<sub>C</sub> alters the frequency dependence of the output voltage divider.
The first objective, to show the lowest pole decreases in frequency, is established using the same approach as the Miller's theorem article. Following the procedure there, Figure 1 is transformed to the electrically equivalent circuit of Figure 2. Application of Kirchhoff's current law to the input side of Figure 2 determines the input voltage to the ideal op amp as a function of the applied signal voltage , namely,
which exhibits a roll-off with frequency beginning at f<sub>1</sub> where
which introduces notation for the time constant of the lowest pole. This frequency is lower than the initial low frequency of the amplifier, which for C<sub>C</sub> = 0 F is .
Turning to the second objective, showing the higher pole increases in frequency, consider the output side of the circuit, which contributes a second factor to the overall gain, and additional frequency dependence. The voltage is determined by the gain of the ideal op amp inside the amplifier as
Using this relation and applying Kirchhoff's current law to the output side of the circuit determines the load voltage as a function of the voltage at the input to the ideal op amp as:
This expression is combined with the gain factor found earlier for the input side of the circuit to obtain the overall gain as
This gain formula appears to show a simple two-pole response with two time constants. It also exhibits a zero in the numerator but, assuming the amplifier gain A<sub>v</sub> is large, this zero is important only at frequencies too high to matter in this discussion, so the numerator can be approximated as unity. However, although the amplifier does have a two-pole behavior, the two time-constants are more complicated than the above expression suggests because the Miller capacitance contains a buried frequency dependence that has no importance at low frequencies, but has considerable effect at high frequencies. That is, assuming the output R-C product, C<sub>L</sub> ( R<sub>o</sub> || R<sub>L</sub> ), corresponds to a frequency well above the low frequency pole, the accurate form of the Miller capacitance must be used, rather than the Miller approximation. According to the article on Miller effect, the Miller capacitance is given by
For a positive Miller capacitance, A<sub>v</sub> is negative. Upon substitution of this result into the gain expression and collecting terms, the gain is rewritten as:
with D<sub>ÃÂ</sub> given by a quadratic in ÃÂ, namely:
Every quadratic has two factors, and this expression simplifies to
where and are combinations of the capacitances and resistances in the formula for D<sub>ÃÂ</sub>. They correspond to the time constants of the two poles of the amplifier. One or the other time constant is the longest; suppose is the longest time constant, corresponding to the lowest pole, and suppose >> . (Good step response requires >> . See Selection of C<sub>C</sub> below.)
At low frequencies near the lowest pole of this amplifier, ordinarily the linear term in ÃÂ is more important than the quadratic term, so the low frequency behavior of D<sub>ÃÂ</sub> is:
where now C<sub>M</sub> is redefined using the Miller approximation as
which is simply the previous Miller capacitance evaluated at low frequencies. On this basis is determined, provided >> . Because C<sub>M</sub> is large, the time constant is much larger than its original value of C<sub>i</sub> ( R<sub>A</sub> || R<sub>i</sub> ).
At high frequencies the quadratic term becomes important. Assuming the above result for is valid, the second time constant, the position of the high frequency pole, is found from the quadratic term in D<sub>ÃÂ</sub> as
Substituting in this expression the quadratic coefficient corresponding to the product along with the estimate for , an estimate for the position of the second pole is found:
and because C<sub>M</sub> is large, it seems is reduced in size from its original value C<sub>L</sub> ( R<sub>o</sub> || R<sub>L</sub> ); that is, the higher pole has moved still higher in frequency because of C<sub>C</sub>.
In short, introducing capacitor C<sub>C</sub> lowered the low pole and raised the high pole, so the term pole splitting seems a good description.
What value is a good choice for C<sub>C</sub>? For general purpose use, traditional design (often called dominant-pole or single-pole compensation) requires the amplifier gain to drop at 20 dB/decade from the corner frequency down to 0 dB gain, or even lower. With this design the amplifier is stable and has near-optimal step response even as a unity gain voltage buffer. A more aggressive technique is two-pole compensation.
The way to position f<sub>2</sub> to obtain the design is shown in Figure 3. At the lowest pole f<sub>1</sub>, the Bode gain plot breaks slope to fall at 20 dB/decade. The aim is to maintain the 20 dB/decade slope all the way down to zero dB, and taking the ratio of the desired drop in gain (in dB) of 20 log<sub>10</sub> A<sub>v</sub> to the required change in frequency (on a log frequency scale) of ( log<sub>10</sub> f<sub>2</sub> − log<sub>10</sub> f<sub>1</sub> ) = log<sub>10</sub> ( f<sub>2</sub> / f<sub>1</sub> ) the slope of the segment between f<sub>1</sub> and f<sub>2</sub> is:
which is 20 dB/decade provided f<sub>2</sub> = A<sub>v</sub> f<sub>1</sub> . If f<sub>2</sub> is not this large, the second break in the Bode plot that occurs at the second pole interrupts the plot before the gain drops to 0 dB with consequent lower stability and degraded step response.
Figure 3 shows that to obtain the correct gain dependence on frequency, the second pole is at least a factor A<sub>v</sub> higher in frequency than the first pole. The gain is reduced a bit by the voltage dividers at the input and output of the amplifier, so with corrections to A<sub>v</sub> for the voltage dividers at input and output the pole-ratio condition for good step response becomes:
Using the approximations for the time constants developed above,
or
which provides a quadratic equation to determine an appropriate value for C<sub>C</sub>. Figure 4 shows an example using this equation. At low values of gain this example amplifier satisfies the pole-ratio condition without compensation (that is, in Figure 4 the compensation capacitor C<sub>C</sub> is small at low gain), but as gain increases, a compensation capacitance rapidly becomes necessary (that is, in Figure 4 the compensation capacitor C<sub>C</sub> increases rapidly with gain) because the necessary pole ratio increases with gain. For still larger gain, the necessary C<sub>C</sub> drops with increasing gain because the Miller amplification of C<sub>C</sub>, which increases with gain (see the Miller equation), allows a smaller value for C<sub>C</sub>.
To provide more safety margin for design uncertainties, often A<sub>v</sub> is increased to two or three times A<sub>v</sub> on the right side of this equation. See Sansen or Huijsing and article on step response.
The above is a small-signal analysis. However, when large signals are used, the need to charge and discharge the compensation capacitor adversely affects the amplifier slew rate; in particular, the response to an input ramp signal is limited by the need to charge C<sub>C</sub>.